Method And Apparatus For Frequency Tracking In A Space Time Transmit Diversity Receiver

ABSTRACT

A system and method for obtaining a frequency error estimate representing the difference between a reference frequency and the frequency of a space-time transmit diversity signal is disclosed. The method includes taking the correlation of total sums, comprised of partial sums taken in defined first and second intervals, to represent the frequency error as the imaginary component of the correlation function.

CROSS-REFERENCE TO RELATED APPLICATIONS

This is a continuation of U.S. application Ser. No. 12/693,591, filedJan. 26, 2010, which is a continuation of U.S. application Ser. No.11/776,884, filed Jul. 12, 2007 (now U.S. Pat. No. 7,676,008), which isa continuation of U.S. application Ser. No. 10/091,772, filed Mar. 6,2002 (now U.S. Pat. No. 7,257,179), which claims priority from U.S.Provisional Application No. 60/273,708, filed Mar. 6, 2001, all theabove applications hereby incorporated herein by reference.

BACKGROUND

1. Field of the Invention

This invention relates to the art of receiving a Space Time TransmitDiversity (STTD) signal. In particular, this invention relates tofrequency tracking of an STTD signal. The invention finds application ina closed-loop automatic frequency control in wireless user equipment.The invention is particularly well suited for use in Personal DigitalAssistants, mobile communication devices, cellular phones, and wirelesstwo-way e-mail communication devices (collectively referred to herein as“wireless devices”). The invention provides utility, however, in anydevice that receives an STTD signal.

2. Description of the Related Art

Space Time Transmit Diversity (STTD) reception is often mandatory foruser equipment (UE), such as mobile communication devices, to operate ina standard fashion with various wireless communication radio networksub-systems (RNS), such as base stations. For example, in the 3rdGeneration Partnership Project (3GPP) standard document No. 3GTS 25.211V3.1.1 (1999-December), it is clearly indicated that STTD reception ismandatory for UE.

The concept of STTD transmission is known to those of skill in the artand involves the use of two transmit antennas at the RNS employing aspace time block coding, such as the example illustrated in the blockdiagram of an STTD encoder of FIG. 1.

Although STTD transmission at an RNS is meant to be beneficial toreception at the UE, frequency tracking at the UE is complicated by STTDtransmission.

Typically, in non-STTD systems, UE tracks an RNS pilot signal in orderto control a local reference oscillator. The pilot signal is usuallyspecifically designed in order to facilitate determining a frequencyoffset.

However, when the received signal from the base station is an STTDsignal, detection of the frequency offset from the received signal ismore difficult. FIG. 2 illustrates a typical pilot modulation patterntransmitted by STTD. The symbol A is a complex number with real andimaginary parts. In this particular case, the symbols from antenna 1 arealways +A, while the symbols from antenna 2 are alternatively +A and −Awith pattern shown. One problem with this pattern is that the two signalcomponents can interfere with each other at the UE. Although notexplicitly illustrated, one of skill in the art will appreciate thatother patterns exist which present the same problem. Considering typicalpropagation conditions between RNS and UE, conventional methods forfrequency tracking do not have sufficient performance to enable reliablefrequency tracking An alternative to frequency tracking is to rely onhighly stable frequency reference source in the UE. However, thisalternative is neither cost effective nor is it optimal from thereceiver performance perspective.

There is a need for a method and apparatus for detecting frequency errorbetween a frequency reference and a received STTD signal at UE. There isa further need for a method and apparatus that controls the frequencyreference by tracking a received STTD signal at UE.

SUMMARY

It is an object of the present invention to obviate or mitigate at leastone disadvantage of previous frequency discriminators for STTD signals

It is a particular object of the present invention to provide a methodand apparatus for detecting frequency error between a frequencyreference and a received STTD signal at UE. It is a further object ofthe present invention to provide a method and apparatus that controlsthe frequency reference by tracking a received STTD signal at UE.

This invention uses the statistical properties of symbols transmitted inan STTD signal to efficiently remove the interference introduced betweenthe two STTD antennas at the UE. Removing the interference provides awide range for frequency error detection, which increases the controlrange, relaxes the requirement for frequency reference accuracy, andeventually reduces UE cost.

In a first aspect, the present invention provides a method of obtaininga frequency error estimate of the difference between a referencefrequency and the frequency of a space time transmit diversity signalfrom first and second received sequences of symbols, transmittedrespectively by first and second antennae, where each sequence has twosets of first and second intervals, such that the contents of the secondinterval of the second received sequence are the additive inverse of thecontents of the first interval of the second received sequence, themethod comprising the steps of: receiving the first and second sequencesof symbols; calculating two sets of first and second partial sums as thesum of the contents of the first and second intervals, respectively, foreach set; calculating total sum functions for the first and second setsby summing the first and second partial sums for each set; calculating acorrelation function based on the total sum functions for the first andsecond sets; and extracting the frequency error estimate from thecorrelation function.

In an embodiment of the first aspect of the present invention thecorrelation function is calculated as a time average of the product ofthe first total sum function and the conjugate of the second total sumfunction. In other embodiments the received symbols are represented bycomplex numbers, and the step of extracting includes isolating theimaginary part of the correlation function as the frequency errorestimate.

The first and second intervals in each set can be adjacent, or they canbe interleaved with the first and second intervals of the other set. Theintervals can also be half or whole symbols in length.

In one embodiment, the step of calculating the total sum includesmultiplying the second partial sum for each set by −1, either inaddition to, or as a replacement of the original total sum step. In afurther embodiment the correlation of the two total sums are added tocreate a third correlation function from which the error can beextracted. In yet another embodiment there is included the step ofmultiplying the frequency error estimate by the average of asignal-to-noise-ratio of the received sequences.

In another embodiment a method of controlling the reference frequency tomatch the frequency of the STTD signal is also provided, using the abovedescribed steps, and further comprising the step of altering thereference frequency based on the frequency error estimate to minimizethe difference between the reference frequency and the frequency of thespace time transmit diversity signal.

In a second aspect of the present invention there is provided anapparatus having a frequency discriminator for obtaining a frequencyerror estimate of the difference between a reference frequency and thefrequency of a space time transmit diversity signal from first andsecond received sequences transmitted respectively by first and secondantennae, and received by a receiving antenna, where each sequence hastwo sets of first and second intervals, of equal length, such that thecontents of the second interval of the second received sequence are theadditive inverse of the contents of the first interval of the secondreceived sequence, the frequency discriminator comprising: a memory,operatively attached to the receiving antenna for storing the contentsof the first and second sequences; interval defining means, operativelyattached to the memory to receive the first and second sequences ofsymbols, for dividing the received sequences into sets of first andsecond intervals; partial sum adding means, operatively attached to theinterval defining means to receive the contents of first and secondsequences during the two sets of first and second intervals, forcalculating two sets of first and second partial sums as the sum of thecontents of the first and second intervals respectively for each set;total sum adding means, operatively attached to the partial sum addingmeans to receive the two sets of first and second partial sums, thetotal sum adding means for calculating total sum functions for the firstand second sets representing the sum of the first and second partialsums for each set; conjugation means, operatively attached to the totalsum adding means to receive the total sum of the second set of partialsums, for calculating the conjugate of the received total sum;multiplier means, operatively attached to the conjugation means andtotal sum adding means to receive the total sums for multiplying thereceived total sums thereby providing a correlation function; and afrequency error estimator, operatively attached to the multiplier meansto receive the correlation function, for extracting the frequency errorfrom the correlation function.

In one embodiment the interval defining means is a sampler andadditionally there is a selective sampler connecting the partial sumadding means and the total sum adding means for selectively providingthe total sum adding means with the partial sum adding means. In otherembodiments the scaling means include means to dividing each total sumby its magnitude, and are ideal scalers. In another embodiment there isa second scaling means, connecting the multiplier means to the diversitycombining means, to receive the multiplied total sums, for scaling thereceived multiplied total sums, and providing the scaled multipliedtotal sums to the diversity combining means.

In other embodiments the frequency error estimator includes a splitterfor separating the real and imaginary component of the correlationfunction to provide the imaginary component of the correlation functionas the frequency error.

The interval defining means include the partial sum adding means withsymbols from adjacent first and second intervals in the same set, oralternatively with symbols from interleaved sets of first and secondintervals. In each of these cases intervals can be one symbol in length,or they can be a half symbol in length.

In other embodiments, the above frequency discriminator includes anegator, that connects the partial sum adding means to the total sumadding means, for multiplying the second partial sum of each set by −1,and provides the negated second partial sum to the total sum addingmeans.

In another embodiment, the frequency error is calculated using anegator, connecting the partial sum adding means to a second total sumadding means, for receiving the second partial sum of each set from thepartial sum adding means, for negating the second partial sum of eachset my multiplying the second partial sum by −1 and a second total sumadding means, operatively attached to the partial sum adding means toreceive the first partial sum of each set and to the negator forreceiving the negated second partial sum for each set, for calculatingsecond total sum functions for the first and second sets representingthe sum of the first partial sum and the negated second partial sum ofeach set, and for providing the conjugation means with the second totalsum of the second set for conjugation. In another embodiment thefrequency error estimator is operatively connected to the diversitycombining means to receive two correlation functions corresponding tothe output of the first and second total sum adding means, for provingthe sum of the two correlation functions as the frequency error.

Further embodiments of this aspect of the invention change the referencefrequency to minimize the frequency error using a loop filter,operatively attached to the frequency discriminator to receive thefrequency error, for generating an oscillator control signal based onthe frequency error to minimize the difference between the referencefrequency and the frequency of the space time transmit diversity signaland a controlled oscillator, operatively attached to the loop filter toreceive the oscillator control signal, for generating the referencefrequency based on the oscillator control signal. In further embodimentsthe controlled oscillator is a numerically controlled oscillator or avoltage-controlled oscillator.

Other aspects and features of the present invention will become apparentto those ordinarily skilled in the art upon review of the followingdescription of specific embodiments of the invention in conjunction withthe accompanying figures.

BRIEF DESCRIPTION OF THE DRAWINGS

Embodiments of the present invention will now be described, by way ofexample only, with reference to the attached Figures, wherein:

FIG. 1 is a prior art block diagram of an STTD encoder;

FIG. 2 is a prior art modulation pattern for an STTD signal;

FIG. 3 is a block diagram that illustrates an AFC at UE;

FIG. 4 is a block diagram that illustrates one embodiment of the methodof operating the FD in an AFC at UE;

FIG. 5 is a block diagram that illustrates one embodiment of the FDapparatus operated on by the method of FIG. 4;

FIG. 6 illustrates an S-curve for the error signal provided by themethod and apparatus of FIGS. 4 and 5;

FIG. 7 is a block diagram that illustrates one embodiment of the methodof operating the FD in an AFC at UE;

FIG. 8 is a block diagram that illustrates one embodiment of the FDapparatus operated on by the method of FIG. 7;

FIG. 9 illustrates an S-curve for the error signal provided by themethod and apparatus of FIGS. 7 and 8;

FIG. 10 is a block diagram that illustrates one embodiment of the methodof operating the FD in an AFC at UE;

FIG. 11 is a block diagram that illustrates one embodiment of the FDapparatus operated on by the method of FIG. 10;

FIG. 12 illustrates an S-curve for the error signal provided by themethod and apparatus of FIGS. 10 and 11;

FIG. 13 is a block diagram that illustrates one embodiment of additionalsteps in the method of operating the FD in an AFC at UE; and

FIG. 14 is a block diagram that illustrates one embodiment of theadditional FD apparatus operated on by the method of FIG. 13.

DETAILED DESCRIPTION

Generally, the present invention provides a method and system fordetermining the frequency error between a reference frequency and thefrequency of an STTD signal. Further embodiments of the inventionprovide a method and system for minimizing the frequency error. Due tothe limited frequency accuracy of the frequency reference typically usedin UE, closed-loop automatic frequency control (AFC) is desired.

FIG. 3 illustrates a block diagram of an AFC loop. The referencefrequency is a generated by a controlled oscillator (CO) 10, such as avoltage or numerically controlled oscillator. The frequencydiscriminator (FD) 20 detects the magnitude and sign of a frequencyerror that the reference frequency may have with respect to a receivedsignal. This frequency error is represented as error signal, e(t), 30and is based on the frequency offset between the frequency reference andthe received signal. The frequency error is then filtered by loop filter(LF) 40 to produce a correction signal applied to the CO to compensatefor the error. In the receiver, after despreading the channel, thecomponent received by antenna 1 is typically a phase-shifted stream ofA's. The component received by antenna 2 is an independently phaseshifted version of a stream of a “+A−A” pattern. If the frequency erroris non-zero, the symbols received by antenna 1 and antenna 2 arestatistically frequency shifted, i.e. a rotation in one direction on acomplex plane. This frequency rotation can be detected by a correlationof the samples with time difference T.

A first embodiment of a method and apparatus for frequencydiscrimination will be described in reference to FIGS. 4-6.

In reference to FIG. 4, one embodiment of a method of frequencydiscrimination to produce a frequency error is illustrated. A first andsecond sequence of symbols, representing the STTD signal, is received,and a first and second intervals are defined in the stream. The secondinterval 250 is defined using the properties of the second sequence,which corresponds to the message transmitted by antenna 2. The secondinterval 250 is defined as the interval where the symbols received arethe same in magnitude as the symbols received in the first interval 240,but differ in their sign. A first partial sum 220A is taken as the sumof the symbols in the two sequences during the first interval 240A, anda second partial sum 230A is taken as the sum of the symbols in the twosequences during the second interval 250A. Upon calculating the partialsums, a total sum, referred to as p(t) 210A, is calculated by adding thetwo partial sums. A second total sum, also referred to as a delayedsignal, is calculated in the same manner during a second interval, andis represented by p(t−τ) 210B, where the time difference between thesets of intervals is τ 260. All references in FIG. 4 referring to thesecond set of intervals are denoted using the same numerals as those forthe first set, but are appended by the letter ‘B’ instead of ‘A’.

This method of computing the total sum 210 allows the properties of theantenna 2 sequence to statistically cancel the interference that theantenna 2 signal would have had on the antenna 1 signal.

As mentioned earlier, the delayed signal p(t−τ) 210B is calculated in amanner analogous to signal p(t) 210A at one time period τ 260B prior totime t. Note that FIG. 4 illustrates the invention by way of exampleonly. As such, in the case of delayed signal p(t−τ) 210B the secondinterval 230B occurs chronologically after the first interval 220B,whereas in the case of signal p(t) 210A, the second interval 230A occurschronologically before the first interval 220A. A person skilled in theart can appreciate that the precise number or order of intervals canvary, as it is dependent on the actual antenna 1 and antenna 2 symbolsequences used, and that the invention can readily be adapted to manysuch symbol sequences, although not expressly shown in the drawings.

After obtaining p(t) and p(t−τ), a correlation of the two functions istaken. In a presently preferred embodiment the correlation is calculatedby taking an average over time of p(t)p*(t−τ), where p*(t−τ) is theconjugate of p(t−τ) as will be understood by one of skill in the art.One of skill in the art will readily appreciated that the frequencyerror can be calculated in a number of ways, and that a presentlypreferred embodiment is to take the imaginary component of the complexnumber representation of the correlation.

Referring to FIG. 5, a multi-finger apparatus 500 that embodies FD 20 ofFIG. 3, employing the above method, can be constructed to provide thefrequency error. The multi-finger structure 500 provides time diversityby having each finger 510 provide a partial correlation which are thencombined by an adder 310 which averages the partial correlations, eachof which is scaled 270, thereby providing diversity combining means. Theoperation of a single finger 510 will be described next.

The received symbols from the two sequences will be stored in a memory205, and will be divided into first and second intervals by an intervaldefining means. A set of adders, for instance found in despreader 207,serving as a partial sum adding means, will add the symbols in each ofthe first and second intervals to provide first and second partial sums.The associated first (selectively available at tap 340A) and second(selectively available at tap 350A) partial sums will then be addedtogether to produce a total sum, the output of adder 210A. This can bein parallel, or in series with the calculation of the time delayedpartial sums (selectively at tap 350B and 340B respectively), which canbe expressed as the partial sums of a second set of intervals. The timedelayed total sum, the output of adder 210B, is provided to a conjugator290, which provides the conjugate of the time delayed total sum. Thefirst total sum, and the time delayed total sum are then scaled byscalers 280A and 280B, which are preferably ideal or exact scalers, andare then multiplied to each other by combiner 300. Scale block 270, toscale the resulting product of the multiplication, is not needed if thescalers 280A and 280B are ideal magnitude or exact normalizers. In apresently preferred embodiment the scale function is defined as

${{scale}(z)} = {\frac{z}{z}.}$

The selective sampler 330 can be designed to sample at some or all theintervals at which the antenna 2 component in the delay line haveopposite sign in the first 340A and second 350A taps, and opposite signin the third 350B and fourth 340B taps. Thus selective sampler 330 onlyprovides symbols to the adders if there are identified first and secondintervals, but the selective sampler can be designed to not provide allsuch instances.

When ideal scalers are used for the second 280A and third 280B “scale”blocks, the detector S-function, an embodiment of which is illustratedin FIG. 6 having chip rate of 3.84 Mcps is defined as:

${e(t)} = {{{Im}\left( \frac{{p(t)} \cdot {p\left( {t - \tau} \right)}^{*}}{{{p(t)}} \cdot {{p\left( {t - \tau} \right)}}} \right)} = {{\sin \left( {\angle \left\lbrack {{p(t)} \cdot {p\left( {t - \tau} \right)}^{*}} \right\rbrack} \right)} = {{\sin \left( {2\pi \; \Delta \; f\; \tau} \right)}.}}}$

Referring to FIG. 6, the control range 32, illustrated by the range ofthe S-function curve, indicates that a frequency error detection rangeof less than 8 kHz is provided.

To increase the control range, a second embodiment of the method andapparatus is provided and will be described in reference to FIGS. 7-9.The block diagram of FIG. 7 illustrates a second embodiment of themethod. This second embodiment changes the correlator by alternating thetap order thereby increasing the control range. The signal p(t) 210A isprovided by the total sum of two partial sums, a first partial sum 220Aand a second partial sum 230A. The first partial sum 220A is the sum ofthe symbols in first interval 240A, while the second partial sum 230A isthe sum of the symbols in second interval 250A. The delay between signalp(t) 210A and p(t−τ) 210B is τ 260, which is half the value of thecorresponding delay 260 in FIG. 4. This shorter delay is due to theinterleaving of the intervals of the two sets. The interleaving of theintervals is done such that the first interval of one set is adjacent tothe first interval of the opposite set, and the second intervals of thetwo sets are adjacent to each other, resulting in a pattern of firstinterval of the first set, second interval of the second set, secondinterval of the first set, and first interval of the second set. Themethod remains the same, save for the interleaving of the intervals.

The corresponding system to this method is illustrated in the apparatusof FIG. 8. The inputs of the adders of the total sum adding means allowthe reorganization of the intervals as described in the method. Theselective sampler 330 samples only at intervals at which the even taps(and odd taps) have opposite sign in the antenna 2 component, to ensurethat the adders are provided only with symbols corresponding to firstand second intervals. The selective sampler 330 can either sample duringall such intervals, or only sample at some of such intervals. TheS-function of this embodiment is illustrated in FIG. 9, which shows thatthe control range 32 has been doubled in this second embodiment andindicates that a frequency error detection range of less than 16 kHz isprovided.

To increase the detection range further, a third embodiment will bedescribed in reference to FIGS. 10-12. The third embodiment uses aspecial spreader 25 that splits the 256-chip pilot symbols into two halfsymbols of 128 chip, so that the first interval 240 and second interval250 are half a symbol, or 128 chip, long. The selective sampler 330samples only during intervals in which the first tap 340A and second tap340B contain half symbols in which the antenna 2 components are oppositein sign to those in respective third tap 350A and fourth tap 350B. Thesampler 330 is restricted to sampling in these intervals, but is notrequired to sample at all these intervals. It should be noted that themethod and system for this embodiment are the same as the previousembodiments, save for the smaller symbol size. This refinement can beemployed in the apparatus and method of either the first or secondembodiments.

Having taught how to eliminate the antenna 2 interference component withrespect to antenna 1 for the purposes of AFC operation, an improvementapplicable to all of the aforementioned embodiments will now bepresented. The improvement makes it possible to independently eliminatethe antenna 1 interference component with respect to antenna 2, therebyproviding a second frequency error signal. The two frequency errorsignals can then combined to provide a third error signal thereby makinguse of the diversity gain provided by an STTD signal.

With some additional steps and apparatus, to be described below inreference to FIGS. 13-14, it is possible to eliminate the antenna 1component in a manner analogous to how the antenna 2 component waseliminated as described in the above embodiments. What will be describedapplies equally well to any of the three embodiments described above,but for the sake of brevity will only be described in reference to thefirst embodiment, as adaptation to the other two would be obvious to aperson skilled in the art.

As compared to the method of FIG. 4, the method of FIG. 13 providesadditional steps to eliminate the antenna 1 component, which can be usedin conjunction with previous methods, or on its own as a separate errordetermining method. Partial sums are taken in each of the first andsecond intervals as in previous embodiments. Where previous embodimentshad summed the partial sums to eliminate the second sequence, thisembodiment takes the additive inverse of the second partial sum, so asto eliminate the first sequence. This can be implemented using eithersubtractors, or a negator, designed to negate a partial sum bymultiplying by −1, in series with an adder. The total sum of the firstpartial sum and the negated second partial sum is q(t) 410A. In ananalogous manner, the delayed signal q(t−τ) 410B is provided. The delaybetween signal q(t) 410A and q(t−τ) 410B is τ 260.

By providing total difference 410 in the invention, statistically theantenna 1 component in the first interval 240 cancels the antenna 1component in the second interval 250 thereby the antenna 1 componentthat would have traditionally interfered with the antenna 2 component inFD 20 operation is eliminated by the invention.

An apparatus adapted to allow the additional steps of eliminating theantenna 1 components is illustrated in FIG. 14. The apparatus positivelysets out the additional hardware required to independently eliminate theantenna 1 component, and although not expressly shown in the drawing ismeant to be operated in conjunction with apparatus that independentlyeliminates the antenna 2 component. Instead of using a first adder 210Aand a second adder 210B as was the case for eliminating antenna 2components, the apparatus for eliminating antenna 1 components uses afirst subtractor 510A and a second subtractor 510B, or as describedabove, it can use a negator on one of the partial sums prior to addingto obtain an additive inverse. By using systems that independentlyeliminate the interference caused by the first and second sequences itis possible to create a third error signal.

Hence while the summation for p(t) 210A and p(t−τ) 210B eliminates thesignal from antenna 2 (assuming infinite channel coherence time), q(t)410A and q(t−τ) 410B eliminates the signal from antenna 1. Bothcorrelation products p(t)p*(t−τ) and q(t)q*(t−τ) are proportional to themagnitude of the carrier-to-interference ratio (CIR) squared andsin(wt). Hence we can just add these together forming an error signalfor the AFC loop of

${e(t)} = {\frac{{Im}\left( {{{p(t)}{p^{*}\left( {t - \tau} \right)}} + {{q(t)}{q^{*}\left( {t - \tau} \right)}}} \right)}{{{{p(t)}{p^{*}\left( {t - \tau} \right)}}} + {{{q(t)}{q^{*}\left( {t - \tau} \right)}}}}.}$

To further illustrate this, let g be the complex channel gain fromantenna 1 to the receiver and let b be the gain from antenna 2. Assumethat b and g are constant for the moment. Then we have

p(t)=2gA(t)e ^(jwt)

q(t)=2bA(t)e ^(jwt),

where “w” is the frequency error. Hence

p(t)p*(t−τ)=4|g| ² A(t)A*(t−τ)e ^(jwt)

q(t)q*(t−τ)=4|b| ² A(t)A*(t−τ)e ^(jwt)

since A(t)=A(t−τ) assuming appropriate despreading, then A(t)A*(t−τ)=1and we have e(t)=sin(wt).

This is the same error signal as in the earlier embodiments. However, alimitation of the previous embodiments is that if the propagation pathfrom the second RNS antenna to the UE is severed then no error signalfor frequency tracking is available. With the improvement outlinedabove, the error signal is always available to the UE unless both firstand second antenna propagation paths from the RNS to the UE are severed,which would result in the loss of all symbols, rendering the loss offrequency error estimates meaningless.

As another enhancement to the above embodiments e(t) can be multipliedby the average signal to noise ratio of the pilot signal, which resultsin a well controlled error signal. Hence, the AFC will have theproperties of a first order Kalman filter that compensates the frequencycontrol strongly when SNR is high and weakly when the SNR is low.

The embodiments of the above-described invention provide three frequencyerror estimates that allow a STTD signal to be tracked in an AFC. Thefirst set of embodiments uses the properties of the second sequence toremove interference that the second sequence causes in the firstsequence, and provides a frequency error estimate based on the firstsequence. The second set of embodiments uses the properties of the firstsequence to remove interference that the first sequence causes in thesecond sequence, and provides a frequency error estimate based on thesecond sequence. A third frequency error estimate is provided throughthe summation of the two previous error estimates, and provides agreater range of frequency error control than either of the first twoindependently.

Additionally, in operation the third error estimate provides additionalrobustness, by maintaining an frequency error calculation in the eventof one transmitting antenna failing. This allows for a level ofredundancy due to the dual sequences transmitted by STTD antennae.

The above-described embodiments of the present invention are intended tobe examples only. Those of skill in the art may effect alterations,modifications and variations to the particular embodiments withoutdeparting from the scope of the invention, which is defined solely bythe claims appended hereto.

1. A frequency discriminator comprising: memory configured to input,from a receiver, first and second sequences of symbols that werewirelessly received by the receiver from respectively first and secondtransmit antennae, wherein each sequence has two sets of first andsecond intervals, and the contents of the second interval of the secondreceived sequence are the additive inverse of the contents of the firstinterval of the second received sequence; a set of adders configured tocalculate two sets of first and second partial sums as the sum of thecontents of the first and second intervals, respectively, for each set;calculate total sum functions for the first and second sets by summingthe first and second partial sums for each set; calculate anauto-correlation function based on the total sum functions for the firstand second sets; and a splitter configured to isolate the imaginary partof the auto-correlation function as a frequency error estimate.
 2. Thefrequency discriminator of claim 1 wherein the auto-correlation functionis calculated as a time average of the product of the first total sumfunction and the conjugate of the second total sum function.
 3. Thefrequency discriminator of claim 1 wherein the auto-correlation functionis calculated as a time average of the product of the first total sumfunction and the conjugate of the second total sum function.
 4. Thefrequency discriminator of claim 1 wherein extracting includes addingthe auto-correlation to a correlation of a second set of total sumfunctions calculated by summing the first partial sum with the additiveinverse of the second partial sum.
 5. The frequency discriminator ofclaim 1 wherein the first and second interval in each set are adjacent.6. The frequency discriminator of claim 1 wherein the first and secondsets of intervals are interleaved with each other.
 7. A frequencydiscriminator comprising: memory configured to input, from a receiver,first and second sequences of symbols that were wirelessly received bythe receiver from respectively first and second transmit antennae,wherein each sequence has two sets of first and second intervals, andthe contents of the second interval of the second received sequence arethe additive inverse of the contents of the first interval of the secondreceived sequence; a set of adders configured to calculate two sets offirst and second partial sums as the sum of the contents of the firstand second intervals, respectively, for each set; multiply the secondpartial sum for each set by −1; calculate total sum functions for thefirst and second sets by summing the first and second partial sums foreach set; calculate an auto-correlation function based on the total sumfunctions for the first and second sets; and a frequency error estimatorconfigured to extract configured to extract the frequency error estimatefrom the auto-correlation function.
 8. The frequency discriminator ofclaim 1 wherein the auto-correlation function is calculated as a timeaverage of the product of the first total sum function and the conjugateof the second total sum function.
 9. The frequency discriminator ofclaim 7 wherein the auto-correlation function is calculated as a timeaverage of the product of the first total sum function and the conjugateof the second total sum function.
 10. The frequency discriminator ofclaim 7 wherein extracting includes adding the auto-correlation to acorrelation of a second set of total sum functions calculated by summingthe first partial sum with the additive inverse of the second partialsum.
 11. The frequency discriminator of claim 7 wherein the first andsecond interval in each set are adjacent.
 12. The frequencydiscriminator of claim 7 wherein the first and second sets of intervalsare interleaved with each other.
 13. A frequency discriminatorcomprising: memory configured to input, from a receiver, first andsecond sequences of symbols that were wirelessly received by thereceiver from respectively first and second transmit antennae, whereineach sequence has two sets of first and second intervals, and thecontents of the second interval of the second received sequence are theadditive inverse of the contents of the first interval of the secondreceived sequence; a set of adders configured to calculate two sets offirst and second partial sums as the sum of the contents of the firstand second intervals, respectively, for each set; calculate total sumfunctions for the first and second sets by summing the first and secondpartial sums for each set; calculate an auto-correlation function basedon the total sum functions for the first and second sets; and afrequency error estimator configured to extract configured to extractthe frequency error estimate from the auto-correlation function. amultiplyer configured to multiply the frequency error estimate by anaverage of a signal-to-noise-ratio of the received sequences.
 14. Thefrequency discriminator of claim 13 wherein the auto-correlationfunction is calculated as a time average of the product of the firsttotal sum function and the conjugate of the second total sum function.15. The frequency discriminator of claim 13 wherein the auto-correlationfunction is calculated as a time average of the product of the firsttotal sum function and the conjugate of the second total sum function.16. The frequency discriminator of claim 13 wherein extracting includesadding the auto-correlation to a correlation of a second set of totalsum functions calculated by summing the first partial sum with theadditive inverse of the second partial sum.
 17. The frequencydiscriminator of claim 13 wherein the first and second interval in eachset are adjacent.
 18. The frequency discriminator of claim 13 whereinthe first and second sets of intervals are interleaved with each other.19. The frequency discriminator of claim 13 wherein the extractingincludes adding the auto-correlation to a correlation of a second set oftotal sum functions calculated by summing the first partial sum with anadditive inverse of the second partial sum.
 20. A communication devicecomprising: the frequency discriminator of claim 13; first and secondreceiving antennas; and the receiver, the receiver being configured togenerate the first sequence from a component received from the firstantenna and to generate the second sequence from a component receivedfrom the second antenna and provide the first and second sequences tothe discriminator.